Device and method for linear listener echo cancellation

ABSTRACT

A device and method utilize a linear feedback/feedforward configuration to estimate and cancel a listener echo wherein equalizer (202) and listener echo cancellation (204 and 208) units are jointly updated, allowing determination of an error signal (204 and 208), in place of the former method of separate error signal determination for each of an equalizer and a listener echo canceller. An improved phase-correction scheme is utilized to compensate the frequency offset in the listener echo. The relationship between the listener echo and talker far echo is utilized to facilitate the listener echo cancellation.

This is a continuation of application Ser. No. 07/791,702, filed on Nov.14, 1991, and now U.S. Pat. No. 5,319,636.

FIELD OF THE INVENTION

This invention relates generally to synchronous digital informationtransmission systems, and more particularly, to echo cancellation insuch transmission systems.

BACKGROUND

Data communication equipment (DCE) such as modems are generally utilizedto transmit and receive data over a communication channel. High speedmodems typically use bandwidth-efficient modulation schemes such asquadrature modulation. In such a system that utilizes binary data, thedata is first mapped into a sequence of complex signal points or symbolsselected from a constellation with a finite number of points. Areal-valued transmitted signal is utilized to carry information aboutthis complex sequence.

One category of DCEs, often referred to as full-duplex modems, iscapable of transmitting and receiving simultaneously over a two-wirecommunication link such as a telephone channel in the general switchedtelephone network. On such a two-wire telephone channel, hybrid couplersare commonly used to perform two to four wire conversions which separatethe data transmission in both directions. Due to imperfect impedancematching at the hybrid couplers, data transmission separations are notideal. Thus, there exist echoes that interfere with normal datatransmission. Such echoes may be divided into two categories: a talkerecho and a listener echo. The talker echo signal, generated by anoutgoing signal from a local modem's transmitter, typically includes anear echo component and a far echo component. The near echo is generatedby hybrids in the local modem and a near-end telephone central office,and the far echo signal is mainly generated by hybrids in a remotecentral telephone office and a remote modem. The far echo signal isdelayed in time relative to the near echo signal since it travels around trip around the telephone channel. The far echo signal may also becorrupted by a frequency offset in the telephone network. The listenerecho signal, in contrast, is generated by a data signal from the remotemodem's transmitter. The overall channel that the listener echo signalpasses through can be viewed as a cascade of the channel that a normalreceived data signal passes through and an extra channel that includesthe round trip telephone trunk and two trans-hybrid paths on both sidesof the trunk, as is illustrated in FIG. 1. The listener echo signal isdelayed in time with respect to the normal received data signal since ittravels an extra round trip around the telephone network. Both the talkfar echo delay and the listener echo delay are about equal to thetelephone trunk round trip delay, and hence are nearly equal to eachother.

Similar to the talker far echo signal corruption, the listener echosignal may also be corrupted by a frequency offset, often designated asphase roll, thereby complicating cancellation of that echo signal. Thus,phase correction circuitry is needed to track and correct phasevariations in the echo.

The listener echo total frequency offset substantially consists of a sumof two components. The first component corresponds to the normalreceived data frequency offset, while the second component correspondsto the frequency offset of the listener echo relative to the normalreceived data signal, which is substantially equal to the talker farecho frequency offset. Further, the second component is substantiallyequal to a sum of the normal data frequency offsets in the transmissionand the reception directions, typically being smaller than the firstcomponent and a total frequency offset.

Most existing two-wire full-duplex high-speed modems come equipped withan adaptive talker echo canceller that is capable of nearly eliminatingthe talker echoes, including the near and the far talker echoes. Thelistener echo is often much weaker than the normal received data signal.The ratio between the normal received data signal and the listener echosignal is substantially computed as the total signal loss by the roundtrip telephone trunk and trans-hybrid losses on both sides of thetelephone trunk. When this ratio is much greater than the requiredsignal-to-noise ratio (SNR), the listener echo signal has little effecton the reception of the remote data. However, as the quality of thetelephone network improves, the signal loss by the telephone trunk maybe reduced, and thus the ratio between the normal received data signaland the listener echo signal may also be reduced. As a result, thelistener echo signal may become a problem, especially for veryhigh-speed modems which may require a higher SNR to obtain reliablereception. In this case, elimination of the listener echo signal becomesnecessary.

In the prior art listener echo cancellers have been implemented as aspecial form decision feedback equalizer. A detected data signal from anoutput of a decision device, after an appropriate delay corresponding toa listener echo delay, is fed back to a transversal filter whosecoefficients are adaptively adjusted such that the filter's output,after correcting a total phase variation caused by a total frequencyoffset in the listener echo signal, becomes an estimate of the listenerecho signal. This estimate is then typically subtracted from aphase-corrected equalizer output before it is sent to the decisiondevice. Such a structure has two potential problems. First, as is thecase for a conventional decision feedback equalizer, decision errors inthe decision device may cause error propagation. Second, to obtain amore reliable decision, complex decision algorithms, such as a Viterbidecoding algorithm, are generally employed. A relatively long decisiondelay may be required for such a decoding algorithm. If the listenerecho delay is shorter than this decision delay, the decision will not bereadily available for the listener echo canceller. In addition, a majordrawback is that a relationship between the listener echo signal and thetalker far echo signal is typically disregarded, forcing estimation ofthe listener echo bulk delay by an extensive search scheme. Prior artcompensates the total frequency offset using a simple first-orderphase-locked loop (PLL). Thus, as is known in the art, an inherentcompensation error by such a first-order PLL is proportional to thefrequency offset to be compensated.

Hence, there is a need for a listener echo signal canceller thateliminates the problem of decision error propagation and that is notlimited by decision delay, that utilizes information obtained for thetalker far echo cancellation, and that improves phase correction.

Summary of the Invention

A device and method are set forth for minimizing a listener echo signalinterference in a substantially demodulated, where desired,communication signal received over a channel having simultaneoustransmission and reception. The device comprises at least a listenerecho canceller for reducing a listener echo interference in a digitalcommunication system having a selected sampling interval, and having asampled input and a sampled output, said sampled input comprisingsubstantially at least a sum of a desired signal and a listener echointerference, and said sampled output being substantially free oflistener echo interference, said listener echo canceller comprising atleast: a bulk delay line means, operably coupled to receive and store atleast one sample of one of: the sampled input and the sampled output ofsaid listener echo canceller, to provide a bulk delay line output; anadaptive listener echo interference estimator means, operably coupled toreceive the bulk delay line output, for generating an estimate of thelistener echo interference; and an interference cancelling means,operably coupled to receive the sampled input of the listener echocanceller and the estimate of the listener echo interference, forsubstantially subtracting said estimate of the listener echointerference from said input of the listener echo canceller.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates how a near echo, a far echo, and a listener echo areformed in a local modem/remote modem system as is known in the art.

FIG. 2A is a block diagram of a first embodiment of a device of thepresent invention.

FIG. 2B is a block diagram of a second embodiment of a device of thepresent invention.

FIG. 2C is a block diagram of a third embodiment of a device of thepresent invention; FIG. 2D and 2E are block diagrams of underlyingexemplary first and second operation modes, respectively, of the thirdembodiment of the device of the present invention; 2D sets forth a blockdiagram of the device of the present invention in an exemplary datareceiving linear feedback mode; 2E sets forth a block diagram of thedevice of the present invention in an exemplary reference directedtraining mode.

FIG. 3 is a block diagram setting forth an exemplary embodiment of thelinear feedback configuration implementation of the device of thepresent invention.

FIG. 4 is a block diagram setting forth an exemplary embodiment of thelinear feedforward configuration implementation of the device of thepresent invention.

FIG. 5 is a block diagram setting forth an exemplary embodiment of thelistener echo estimator of FIGS. 3 and 4.

FIG. 6 is block diagram of a local modem and a remote modem withprecoding as is known in the art.

FIG. 7 is an exemplary embodiment of a device of the present inventionhaving a linear feedback configuration with precoding.

FIG. 8 is an exemplary embodiment of a device of the present inventionhaving a linear feedforward configuration with precoding.

FIG. 9 is a block diagram of setting forth exemplary embodiments of alistener echo phase estimating unit (LEPE), a listener echo phase-lockedloop unit (LEPLL), a listener echo phase correcting unit (LEPC), and alistener echo phase correction inverse unit (LEPC)I of FIGS. 3 and 4.

FIG. 10 is a block diagram setting forth exemplary embodiments of areceived data phase estimating unit (RDPE), a received data phase-lockedloop unit (RDPLL), a received data phase correcting unit (RDPC), and areceived data phase correction inverse unit (RDPCI) of FIGS. 3 and 4.

FIG. 11 is a block diagram setting forth an exemplary embodiment of amodem that includes a listener echo cancelling device in accordance withthe present invention.

DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT

The present invention provides a device and method for an equalizer thatincludes a listener echo cancellation system that is stable, typicallylinear, and has a listener echo cancellation residual error thatconverges to a global minimum. In the present invention, the listenerecho phase correction means needs to compensate only the listener echofrequency offset relative to the received data signal, which istypically smaller than the listener echo total frequency offset andeasier to compensate. The present invention also utilizes a fact thatsuch a frequency offset component is substantially equal to a talker farecho frequency offset, an estimate of which is typically available forfar echo cancellation. In addition, the present invention utilizes afurther fact, that the listener echo bulk delay relative to the receiveddata signal is substantially equal to the talker far echo bulk delayrelative to the near echo that is usually available for far echocancellation. Further, an equalizer and a listener echo canceller arejointly updated to allow determination of an error signal only once,more efficiently than in the past where a separate error signal wasdetermined for each of the equalizer and the listener echo canceller.

FIG. 2A is a block diagram of a first embodiment of the presentinvention, illustrating a feedback implementation of a listener echocanceller (LECR). An interference cancelling unit (201), operablycoupled to receive a sampled input to the listener echo canceller and anestimate of a listener echo interference from an adaptive listener echointerference estimator unit (203), substantially subtracts the estimateof the listener echo interference from the sampled input to provide alistener echo corrected output. The LECR of FIG. 2A further includes abulk delay line unit (205), operably coupled to receive the listenerecho corrected output, for providing a bulk delay line output to thelistener echo interference estimator unit. Further description of atypical listener echo estimator unit and a bulk delay line unit are setforth below.

FIG. 2B is a block diagram of a second embodiment of the presentinvention, illustrating a feedforward implementation of a listener echocanceller (LECR). An interference cancelling unit (211), operablycoupled to receive a sampled input to a listener echo canceller and anestimate of a listener echo interference from an adaptive listener echointerference estimator unit (209), substantially subtracts the estimateof the listener echo interference from the sampled input to provide alistener echo corrected output. The LECR of FIG. 2B further includes abulk delay line unit (207), operably coupled to receive the sampledinput, for providing a bulk delay line output to the listener echointerference estimator unit. Further description of a typical listenerecho estimator unit and a bulk delay line unit are set forth below.

FIG. 2C is a block diagram of a third embodiment of a device of thepresent invention for minimizing a listener echo signal interference ina substantially demodulated, equalized communication signal receivedover a channel having simultaneous transmission and reception. Thedevice comprises at least: an equalizing unit (202), a first correctingunit (204), and a second correcting unit (208). The equalizing unit(202) receives the demodulated, where desired, communication signal andprovides an equalized signal to the first correcting unit (204). Thefirst correcting unit (204), operably coupled to the equalizing unit(202) and to the second correcting unit (208), at least receives theequalized signal from the equalizing unit (202), provides a listenerecho estimate with a proper phase correction utilizing a phase-trackingdevice, corrects the equalized signal in accordance with that listenerecho estimate, and provides a first corrected signal to the the secondcorrecting unit (208). The second correcting unit (208) is operablycoupled to the first correcting unit (204) and is configuredsubstantially for providing a phase correction to the first correctedsignal, outputting a second corrected signal to the decoder, providingan error signal to the equalizing unit (202) for updating the equalizercoefficients and to the first correcting unit (204) for updating thelistener echo estimator coefficients, and, where desired, providing areference signal to the first correcting unit (204) in the referencedirected training mode for training the listener echo estimator and thephase-tracking device.

FIG. 2D and 2E are basic signal-path block diagrams of underlyingexemplary first and second operation modes, respectively, of the thirdembodiment of the device of the present invention. An error signalprovided by a second correcting unit (224) to the equalizer unit (210)and to the first correcting unit (212) for updating the equalizer unit(210) and listener echo estimator coefficients is not illustrated. FIG.2D sets forth a block diagram of the device of the present invention inan exemplary data receiving linear feedback implementation wherein aninput to a listener echo estimator (220) is provided via a linearfeedback configuration. An equalizer unit EQ (210) is coupled to receivea demodulated communication signal. The first correcting unit (212) isoperably coupled to the equalizer unit (210) and to the secondcorrecting unit (224), and comprises a listener echo estimator LEE (220)with a switch S1 (222) for selecting an input signal to the LEE (220) indifferent operation modes, a listener echo phase correction unit LEPC(216) and a combining unit (214) that substantially obtains a differencebetween the equalized demodulated communication signal and thephase-corrected listener echo estimate for providing a first correctedsignal to the second correcting unit (224). The second correcting unit(224) includes a received data phase correction unit (226) to providephase correction to the first corrected signal from the first correctingunit (212) and output the second corrected signal u to a decoder. InFIG. 2D, the device is in a communication data receiving mode. Theswitch S1 (222) selects z, an output of the combining unit (214) as aninput to the LEE (220). Hence, the first correcting unit (212) operatesin a linear feedback mode.

FIG. 2E sets forth a block diagram of the device of the presentinvention in an exemplary linear feedback listener echo cancellingimplementation with a reference-directed training mode. In such anoperation mode, a switch S2 (232) in the second correcting unit (224)selects a reference signal as signal v, the reference signal being apredetermined data signal transmitted by a remote modem in areference-directed training period and being known by a local modemreceiver. Signal v is phase-rotated in a received data phase correctioninverse unit RDPCI (230) that performs an inverse operation of that of areceived data phase correction unit RDPC (226), the RDPCI unit (230)outputting signal d. Signal d is selected by the switch S1 (222) in thefirst correcting unit (212) as an input to the LEE (220).

FIG. 3 is a block diagram setting forth an exemplary embodiment of alinear feedback configuration implementation of the device of thepresent invention. The device minimizes listener echo signalinterference in a substantially demodulated, where desired,communication signal received over a channel having simultaneoustransmission and reception, the device comprising at least: anequalizing unit (302), operably coupled to receive the demodulated,where desired, communication signal, for substantially providing anequalized signal; a first correcting unit (304), operably coupled to theequalizing unit (302) and to a second correcting unit (308), and beingconfigured for at least: receiving the equalized signal; determining atleast a phase-corrected listener echo estimate; and combining theequalized signal with the phase-corrected listener echo estimate toprovide at least a first corrected signal; wherein the second correctingunit (308) is operably coupled to the first correcting unit (304) and tothe equalizing unit (302), and is configured substantially for at least:receiving the first corrected signal, providing a phase correction tothe first corrected signal to generate a second corrected signal that isoutput to a decoder, providing an error signal to the equalizing unit(302) for updating coefficients of the equalizing unit (302) and to thefirst correcting unit (304) for updating coefficients of the LEE (320),and providing, where a reference-directed training mode is selected, areference signal as an input signal to the LEE (320) in the firstcorrecting unit (304).

In the exemplary embodiment of the linear feedback configurationimplementation, FIG. 3, the first correcting unit (304) comprises atleast: a primary combining unit (310), operably coupled to theequalizing unit (302) and to a listener echo phase correcting unit LEPC(316) that provides at least a phase-corrected listener echo estimatee_(L), for substantially obtaining a difference between an equalizedsignal q and the phase-corrected listener echo estimate e_(L), being thefirst corrected signal z; a first switching unit S1 (322), operablycoupled to the primary combining unit (310) and to a second correctingunit (308), for selecting, as desired, one of:

for a data receiving mode, substantially a linear feedback signal z thatis substantially an equalized, listener echo estimate corrected signal,and

for a reference-directed training mode, a phase-rotated reference signald;

a listener echo estimating unit LEE (320) having LEE coefficients, beingoperably coupled to the first switching unit (S1) for utilizing theselected signal to determine a first listener echo estimate and operablycoupled to a listener echo phase correction inverse unit LEPCI (318)that provides a rotated error signal, for utilizing the rotated errorsignal to update the LEE coefficients; wherein the listener echo phasecorrection unit LEPC (316) is operably coupled to the listener echoestimating unit LEE (320) and to a listener echo phase-locked loop unitLEPLL (314) for providing a phase correction to the listener echoestimate and for generating a phase-corrected listener echo estimatee_(L) to the primary combining unit (310) and is responsive to acontrolled input of the LEPLL (314); and a listener echo phase errordetermining unit LEPE (312), operably coupled to the equalizing unit andto the listener echo phase correcting unit LEPC (316), for utilizing atleast the equalized signal q and the phase-corrected listener echoestimate e_(L) to provide a listener echo phase error θ to the LEPLL(314); wherein the listener echo phase-locked loop unit LEPLL (314) isoperably coupled to the listener echo phase error determining unit LEPE(312) for tracking a correct phase of a listener echo based at least onθ and for providing a phase correction Ψ to the LEPC (316) and to theLEPCI (318); and wherein the listener echo phase correction inverse unitLEPCI (318) is operably coupled to the listener echo phase-locked loopunit LEPLL (314) and to the second correcting unit (308) that providesthe error signal ε, for substantially rotating the error signal ε fromthe second correcting unit in accordance with the LEPLL phase correctionΨ to provide a rotated error signal η for updating coefficients of theLEE (320).

In the above embodiment the second correcting unit typically comprisesat least: a received data phase correcting unit RDPC (324), operablycoupled to the first correcting unit (304) and to a received dataphase-locked loop unit RDPLL (330), for substantially providing a datasignal phase correction to the first corrected signal z and utilizingthat data phase correction to generate a second corrected signal u foroutputting to a decoder; a preliminary decision unit (PD) (328),operably coupled to the received data phase correcting unit RDPC (324),for generating an output signal w in accordance with the secondcorrected signal u, w being a preliminary estimate of a data signaltransmitted by a remote modem; a second switching unit (S2), operablycoupled to the preliminary decision unit PD (328) and to a referencesignal unit (REFERENCE) having at least a reference signal, forselecting one of: for a data receiving mode, the preliminary decisionoutput signal w, and for a reference-directed training mode, apre-determined reference signal r, as a decision signal v; a receiveddata phase error unit RDPE (328), operably coupled to the received dataphase correcting unit RDPC (324) and to the second switching unit S2(336), for substantially determining a received data phase error α inaccordance with signals u and v; a received data phase-locked loop unitRDPLL (330), operably coupled to the received data phase correcting unitRDPC (324) and to a received data phase correction inverse unit RDPCI(332), for substantially tracking a phase of a received data signal andproviding a phase correction estimate φ(n) to the RDPC (324) and to theRDPCI (332) in accordance with α; the received data phase correctioninverse unit RDPCI (332) being operably coupled to the received dataphase-locked loop unit RDPLL (330) and to the second switching unit S2(336), for performing an inverse operation of the RDPC to provide aphase rotation on decision signal v that generates a rotated decisionsignal d; and a second combining unit (334), operably coupled to thereceived data phase correction inverse unit and to the first correctingunit, for substantially subtracting the first corrected signal z from dto provide an error signal ε(n) to the equalizing unit (302) forupdating equalizer unit coefficients and to the first correcting unit(304) for updating LEE coefficients after phase rotation in the LEPCI.The second combiner is typically an adder.

FIG. 4 is a block diagram setting forth an exemplary embodiment of alinear feedforward configuration implementation of the device of thepresent invention. The exemplary embodiment in FIG. 4 is similar to thatin FIG. 3, except that the first switching unit S1 (420) selects theequalized signal q instead of the first corrected signal z as the inputto the LEE (422). Thus, in this exemplary embodiment, an equalizing unit(402) is utilized also, and the first correcting unit (404) comprises atleast: a first adding unit (418), operably coupled to the equalizingunit (402) and to a listener echo phase correcting unit LEPC (416) thatprovides at least a phase-corrected listener echo estimate, forsubstantially obtaining a difference between the equalized signal andthat listener echo estimate, being the first corrected signal; aswitching unit S3 (420), operably coupled to the equalizing unit (402)and to the second correcting unit (408), wherein the second correctingunit (408) provides a rotated reference signal, for selecting, asdesired, one of: the equalized signal, and the rotated reference signal,as a selected signal; a listener echo estimating unit LEE (422) havingLEE coefficients, operably coupled to the switching unit S3 (420) forutilizing the selected signal to determine a listener echo estimate, andto a listener echo phase correction inverse unit LEPCI (414) thatprovides a rotated error signal, for utilizing the rotated error signalto update the LEE coefficients; wherein the listener echo phasecorrecting unit LEPC (416) is operably coupled to the listener echoestimating unit LEE (422) and to a listener echo phase-locked loop unitLEPLL (412) that provides a listener echo phase correction estimate, forsubstantially rotating the listener echo estimate of the LEE inaccordance with the listener echo phase correction estimate to determineat least a phase-corrected listener echo estimate; a listener echo phaseerror determining unit LEPE (410), operably coupled to the equalizingunit (402) and to the listener echo phase correcting unit LEPC (416),for utilizing the equalized signal and the listener echo estimate toprovide a listener echo phase error; wherein the listener echophase-locked loop unit LEPLL (412) is operably coupled to the listenerecho phase error determining unit LEPE (410), for utilizing the listenerecho phase error to provide a listener echo phase correction estimate;and wherein the listener echo phase correction inverse unit LEPCI (430)is operably coupled to the listener echo phase-locked loop unit LEPLL(412) and to the second correcting unit (408) that provides an errorsignal, for substantially utilizing the listener echo phase correctionestimate and the second correcting unit error signal for providing arotated error signal for updating the listener echo estimating unit. Thesecond correcting unit (408) typically is substantially that describedabove.

The equalizing unit, having equalizing coefficients, is implementedutilizing known techniques. Typically a baseband adaptive equalizer maybe used where an input is a demodulated baseband signal, as isillustrated in the FIGS. described above. A passband adaptive equalizermay be used where an input is a modulated passband signal. In the lattercase, a demodulator follows the equalizer. A least mean square (LMS)algorithm is typically used for updating the coefficients of theequalizer.

The listener echo estimating unit LEE in FIGS. 3 and 4 (320, 422) isfurther illustrated in FIG. 5. Typically, the LEE includes at least: abulk delay line (BD) (502), operably coupled to the first switching unitS1 (322, 420), for providing a delay substantially in accordance withBL, a listener echo bulk delay relative to a received data signal; andan adaptive transversal filter (504), operably coupled to BD (502),whose coefficients are a_(k) (n), where k=0, 1, 2, . . . , M-1, where Mis a number of taps in the transversal filter unit, and whose output issubstantially ##EQU1## where n is a discrete time index and g() is inputfrom the first switching unit. The delay B_(L), where desired, issubstantially determined by a talker far echo delay measuring method,based on the proposition that, both the listener echo delay and thetalker far echo delay being actually substantially equal to acommunication round trip delay, the said delays are substantially equalto each other.

Devices such as modems may include trellis precoding technique, wheredesired, as illustrated in FIG. 6, wherein a local modem A communicateswith a remote modem B. Thus, in another embodiment, illustrated in FIG.7, of a linear feedback listener echo canceller of the present inventionwherein precoding is utilized, there is further included a noiseprediction filter NPF (702), operably coupled between the equalizingunit (302) and the first correcting unit (304), to receive and filterthe demodulated, equalized received communication signal, so that aspectrum of noise in the demodulated, equalized communication signal iswhitened; and an inverse noise prediction filter NPF⁻¹ (704), operablycoupled between the second correcting unit (308) and the equalizing unit(302), for filtering an error signal ε(n) of the second correcting unitto provide an adjusted error for updating equalizer coefficients.

Similarly, in an embodiment, illustrated in FIG. 8, of a linearfeedforward listener echo canceller of the present invention wheretrellis precoding is utilized, a noise prediction filter NPF (802) andan inverse noise prediction filter NPF⁻¹ (804) are included in a fashionsimilar to that described above for FIG. 7.

The listener echo cancelling device of the present invention istypically implemented in a modem utilized for communicating with aremote device in both directions over a channel, described as above,

Exemplary determinations described above are more particularly set forthbelow, wherein the following terminology is utilized: an equalizedsignal is q(n), n being the discrete time index; a rotated error signalis η(n); a phase-corrected listener echo estimate is e_(L) (n); a firstcorrected signal is z(n); a first listener echo estimate is p(n); and anerror signal is ε(n); a received communication signal phase error isα(n); a received communication signal phase correction estimate is φ(n);a remote transmitted communication signal estimate is v(n), being oneof: a known reference signal r(n) for selection of a reference-directedtraining mode, and a preliminary decision of the remote transmittedcommunication signal communication signal w(n); a phase-rotated decisionsignal is d(n);

The phase-corrected listener echo estimate is substantially of a form:e_(L) (n)=p(n)e^(j) Ψ(n), where n is a discrete time index, p(n) is afirst listener echo estimate, and Ψ(n) is a listener echo phasecorrection estimate. The first corrected signal z(n) is substantiallydetermined as a difference between q(n) and e_(L) (n): z(n)=q(n)-e_(L)(n), where q(n) is the equalized signal.

An exemplary embodiment of a phase correcting device in the firstcorrecting unit is set forth in FIG. 9, wherein the units LEPE (902),LEPLL (904) and LEPCI (906) are included. The LEPE (920) determines alistener echo phase error θ(n), a phase difference between the equalizedsignal q(n) and the phase-corrected listener echo estimate e_(L) (n),utilizing a typical method of determining phase differences between twosignals as is known in the art (for example, as in U.S. Pat. No.4,813,073 and in U.S. Pat. No. 4,987,569): ##EQU2## where e_(Lr) ande_(Li) are the real and imaginary parts, respectively, of e_(L) (n), andq_(r) and q_(i) are the real and imaginary parts, respectively, of theequalized signal q(n).

The exemplary embodiment of the LEPE (902), FIG. 9, includes a firstdevice (910) that provides a real part q_(r) (n), an imaginary partq_(i) (n), and a magnitude of q(n) and a second device (912) thatprovides a real part e_(Lr) (n), an imaginary part e_(Li) (n), and amagnitude of e_(L) (n), first and second multipliers (914, 916) operablycoupled to the first and second devices, for providing the productse_(Li) (n)q_(r) (n) and e_(Lr) (n)q_(i) (n), a first adder (920)operably coupled to the first and second multipliers substantiallydetermines a difference e_(Li) (n)q_(r) (n)-e_(Lr) (n)q_(i) (n), a thirdmultiplier (918) operably coupled to the first device and to the seconddevice provides |q(n)||e_(L) (n)|, and a first divider (922) providesθ(n).

With the configuration in the present invention, the LEPC unitcompensates only the listener echo frequency offset relative to thereceived data signal, which is generally much smaller than the totalfrequency offset, making compensation easier to achieve.

In this embodiment the listener echo phase locked loop LEPLL (980) isimplemented as a typical second order digital phase locked loop as isknown in the art, for example, as described in U.S. Pat. No. 4,813,073and U.S. Pat. No. 4,987,569, wherein a listener echo frequency offsetrelative to the received data signal f_(L) and a listener echocorrection Ψ(n) are determined recursively utilizing the listener echophase error θ(n) obtained in the LEPE:

a phase-locked loop of the first correcting unit determines an estimateof f_(L) and a phase correction estimate for p(n+1), wherein,

    f.sub.L (n+1)=f.sub.L (n)+b.sub.1 θ(n),

and

    Ψ(n+1)=Ψ(n)+b.sub.2 θ(n)+b.sub.3 f.sub.L (n+1),

where b₁, b₂, and b₃ are constants that determine a phase-locked loopcharacteristic.

Based on the proposition that f_(L) is substantially equal to a talkerfar echo frequency offset f_(F), as soon as an estimate of f_(F) isavailable, for example, using the method described in U.S. Pat. No.4,987,569, that estimate of f_(F) may be utilized to initialize anestimate of f_(L) in the LEPLL. This initialization scheme facilitatesan initial convergence of the LEPLL.

In the exemplary embodiment illustrated in FIG. 9, the listener echophase-locked loop LEPLL (980) includes at least a third multiplier(928), operably coupled to first divider (922) of the LEPE (902) thatmultiplies the listener echo phase error θ(n) by a first constant b₁ toprovide b₁ θ(n), a fourth multiplier (93S), coupled to the first divider(922) of the LEPE (902), that multiplies θ(n) by a second constant b₂ toprovide b₂ θ(n), a second adder (930), operably coupled to the thirdmultiplier (928) and to a first delay unit (932), that substantiallyadds b₁ θ(n) to a current estimate of f_(L) to provide an updatedestimate of f_(L) which is then stored in the first delay unit (932),and is multiplied by a third constant b₃ utilizing a fifth multiplier(934) that is operably coupled to the second adder (930) to provide afifth multiplier output. A third adder (938), operably coupled toreceive the fifth multiplier output and b₂ θ(n) is utilized for addingsame to b₂ θ(n) to provide a third adder output. A fourth adder (940),operably coupled to receive the third adder output and a second delayunit (942) output, is utilized to add the third adder output to acurrent value of Ψ(n) to generate an updated value of Ψ(n) that isstored in the second delay unit (942) and is further transmitted to theLEPC (904) and to the LEPCI (906). The second delay unit is operablycoupled to provide the current value of Ψ(n) to the fourth adder (940).

A second approach (not shown) may also be used to implement the LEPLL.Ψ(n) is determined based on the sign of q(n), s(n):

    s(n)=sign[q(n)]=sign [e.sub.Li (n)q.sub.r (n)-e.sub.Lr (n)q.sub.i (n)],

defining

    s(n)=1 for e.sub.Li (n)q.sub.r (n)-e.sub.Lr (n)q.sub.i (n)>TH,

    s(n)=-1 for e.sub.Li (n)q.sub.r (n)-e.sub.Lr (n)q.sub.i (n)<-TH,

and

    s(n)=0;

otherwise, where TH is a small predetermined positive threshold value;and determining

    f.sub.L (n+1)=f.sub.L (n)+d.sub.2 s(n),

and

    Ψ(n+1)=Ψ(n)+d.sub.2 s(n)+b.sub.3 f.sub.L (n+1),

where d₁ and d₂ are selected constants and f_(L) (n), where desired, isinitialized using a talker far echo frequency offset estimate tofacilitate an initial convergence. The second approach iscomputationally more efficient than the first approach. An initialconvergence utilizing the second approach is facilitated by anappropriate initialization scheme using the talker far echo frequencyoffset estimate.

An exemplary embodiment of the LEPC (904), utilized to determine e_(L)(n), is set forth in FIG. 9, wherein, upon input of Ψ(n) from the LEPLL(908), a first unit (1ST EXP) (924) for providing powers of e providese^(j) Ψ(n) to a sixth multiplier (925) that is operably coupled to the1ST EXP (924) and to receive p(n), for substantially multiplying e^(j)Ψ(n) and p(n) to obtain e_(L) (n): e_(L) (n)-p(n)e^(j) Ψ(n).

An exemplary embodiment of the LEPCI (906), utilized to determine η(n),is set forth in FIG. 9, wherein, upon input of Ψ(n) from the LEPLL(908), a second unit (2ND EXP) (926) for providing powers of e providese-^(j) Ψ(n) to a seventh multiplier (927) that is operably coupled tothe 2ND EXP (924) and to receive ε(n), for substantially multiplyinge-^(j) Ψ(n) and ε(n) to obtain η(n): η(n)=ε(n)e-^(j) Ψ(n).

η(n) is utilized to update the listener echo estimator (LEE)coefficients, typically according to an LMS-type algorithm:

    a.sub.k (n+1)=a.sub.k (n)+δη(n) g(n+B.sub.L +k), k=0,1,. . . , M-1,

where δ is a small adaptation constant and M is a number of taps in theLEE.

Received data phase correction may be implemented using known phasecorrection techniques (for example, as described in U.S. Pat. No.4,813,073). An exemplary embodiment of a received data phase errordetermining unit (RDPE) (1002), a received data phase-locked loop(RDPLL) (1008), a received data phase correction unit (RDPC)(1004), anda received data phase correction inverter unit (RDPCI) (1006) of FIGS. 3and 4 is set forth in FIG. 10. The RDPE (1002) is utilized to determineda phase error α(n) using signals v(n) and u(n) such that: ##EQU3## wherev_(r) (n) and v_(i) (n) are real and imaginary parts, respectively, ofv(n), and u_(r) (n) and u_(i) (n) are real and imaginary parts,respectively, of u(n). A third device (1010) provides a real part v_(r)(n), an imaginary part v_(i) (n), and a magnitude of v(n), and a seconddevice (1012) provides a real part u_(r) (n), an imaginary part u_(i)(n), and a magnitude of u(n), eighth and ninth multipliers (1014, 1016)operably coupled to the first and second devices, provide the productsu_(i) (n)v_(r) (n) and u_(r) (n)v_(i) (n), a fifth adder (1018),operably coupled to the eighth and ninth multipliers, substantiallydetermines a difference u_(i) (n)v_(r) (n)-u_(r) (n)v_(i) (n), a squareunit (1020) operably coupled to the third device, provides |v(n)|², anda second divider (1022) operably coupled to the square unit (1020)provides α(n). Alternatively, a prestored table of 1/|v(n)|² may beemployed, allowing replacement of the second divider (1022) by a look-uptable together with an associated multiplier.

In the above embodiment, the received data phase-locked loop RDPLL(1008) is implemented as a typical second-order digital phase-lockedloop as, for example, in U.S. Pat. No. 4,813,073, wherein an estimate ofa received data frequency offset f_(D) and a received data phasecorrection φ(n) are determined recursively utilizing the received dataphase error α(n) obtained in the RDPE (1002), substantially as set forthbelow:

    f.sub.D (n+1)=f.sub.D (n)+c.sub.1 α(n);

such that

    φ(n+1)=φ(n)+c.sub.2 α(n)+c.sub.3 f.sub.D (n+1) ,

where c₁, c₂, and c₃ are constants that determine a desired phase-lockedloop characteristic.

In the exemplary embodiment, FIG. 10, a received data phase errordetermining unit RDPE (1002), a received data phase-locked loop RDPLL(1008), a received data phase correcting unit RDPC (1004), and an RDPCI(1006) of FIGS. 3 and 4 are included. The RDPLL (1008) includes at leasta tenth multiplier (1028), operably coupled to a second divider (1022)of the RDPE (1002) that multiplies the phase error α(n) by a fourthconstant c₁ to provide c₁ α(n), an eleventh multiplier (1036), coupledto the second divider (1022) of the RDPE (1002), that multiplies α(n) bya fifth constant c₂ to provide c₂ α(n), a sixth adder (1030), operablycoupled to the tenth multiplier (1028) and to a third delay unit (1032),that substantially adds c₁ α(n) to a current estimate of f_(D) toprovide an updated estimate of f_(D) which is then stored in the thirddelay unit (1032), and is multiplied by a sixth constant c₃ utilizing atwelfth multiplier (1034) that is operably coupled to the sixth adder(1030) to provide a twelfth multiplier output. A seventh adder (1038),operably coupled to receive the twelfth multiplier output and c₂ α(n),is utilized for adding same to c₂ α(n) to provide a seventh adderoutput. A eighth adder (1040), operably coupled to receive the seventhadder output and a fourth delay unit (1042) output, is utilized to addthe seventh adder (1038) output to a current value of φ(n) to generatean updated value of φ(n) that is stored in the fourth delay unit (1042)and is further transmitted to the RDPC (1004) and to the RDPCI (1006).The fourth delay unit (1042) is operably coupled to provide the currentvalue of φ(n) to the eighth adder (1040).

An exemplary embodiment of the RDPC (1004), utilized to determine u(n)isset forth in FIG. 10, wherein, for an input of φ(n), a third unit 3RDEXP (1024) for providing powers of e provides e^(j)φ (n) and athirteenth multiplier (1025), operably coupled to the third unit 3RD EXP(1024) and to receive an input z(n), substantially multiplies e^(j)φ(n)times z(n) to obtain u(n): u(n)=z(n)e^(j)φ (n).

An exemplary embodiment of the RDPCI (1006), utilized to determine d(n)is set forth in FIG. 10, wherein, for an input of φ(n), a fourth unit4TH EXP (1026) for providing powers of e provides e-^(j) φ(n)and afourteenth multiplier (1027), operably coupled to the fourth unit 4THEXP (1026) and to receive v(n), substantially multiplies e-^(j) φ(n)times v(n) to obtain d(n):

d(n)=v(n)e-^(j) φ(n).

d(n) is utilized to determine the error signal

    ε(n)=d(n)-z(n),

which in turn is used to update the equalizer coefficients, typicallyaccording to the LMS algorithm. d(n) is also sent to the firstcorrecting unit for updating the LEE coefficients after phase rotationin the LEPCI.

The method of the present invention (not illustrated) for minimizing alistener echo signal interference in a substantially demodulated, wheredesired, equalized communication signal received over a channel havingsimultaneous transmission and reception, the method comprising at leastthe steps of: receiving the demodulated, where desired, communicationsignal, for substantially providing an equalized signal utilizing anequalizing unit; determining at least a first listener echo estimate;providing an appropriate phase correction to the first listener echoestimate in a listener echo estimating unit; and combining the equalizedsignal with the phase-corrected listener echo estimate to provide atleast a first corrected signal; selecting an appropriate signal in adifferent operational mode for the listener echo estimation; providingat least an error signal for updating the equalizing unit and listenerecho estimating unit coefficients; providing a phase correction to thefirst corrected signal to generate a second corrected signal, andoutputting substantially the second corrected signal, the secondcorrected signal being a listener echo corrected and phase correctedcommunication signal.

The method typically further includes one of the following twoapproaches:

including at least the steps of: substantially obtaining a differencebetween the equalized signal and the phase-corrected listener echoestimate, being the first corrected signal; selecting, as desired, oneof: the first corrected signal, and a phase-rotated reference signal, asa first selected signal; utilizing the first selected signal todetermine a first listener echo estimate; providing a phase correctionto the listener echo estimate and substantially obtaining aphase-corrected listener echo estimate; utilizing an equalized signaland the phase-corrected listener echo estimate to provide a listenerecho phase error; utilizing the listener echo phase error to provide alistener echo phase correction estimate; providing an error signal andsubstantially rotating the error signal based on the listener echo phasecorrection estimate to obtain a rotated error signal for updating thelistener echo estimation; and

including at least the steps of: providing at least the phase-correctedlistener echo estimate and substantially obtaining a difference betweenthe equalized signal and the phase-corrected listener echo estimate,being the first corrected signal; selecting, as desired, one of: theequalized signal, and a phase-rotated reference signal, as a selectedsignal; utilizing this second selected signal to determine a firstlistener echo estimate; providing to the first listener echo estimate aphase correction and substantially obtaining a phase-corrected listenerecho estimate; utilizing the equalized signal and the phase-correctedlistener echo estimate to provide a listener echo phase error; utilizingthe listener echo phase error to provide a listener echo phasecorrection estimate; providing an error signal and substantiallyrotating the error signal based on the listener echo phase correctionestimate to obtain a rotated error signal for updating the listener echoestimate.

In addition to utilization of one of the two approaches set forth above,the method of the present invention typically further includes at leastthe steps of: substantially determining a data phase correction andutilizing that data phase correction to provide a second correctedsignal, being substantially a listener echo corrected, phase correctedcommunication signal; providing a preliminary decision signal as apreliminary estimate of the data signal transmitted by a remote modem;selecting, as desired, one of: the preliminary decision signal and aknown reference signal as a second selected signal; substantiallyproviding a received data phase correction estimate; providing aphase-rotation to the second selected signal to obtain a rotateddecision signal; and substantially subtracting the first correctedsignal from the rotated decision signal to obtain an error signal forequalization unit and listener echo estimating unit adaptation.

Also, the method of the present invention generally includes at theleast the following steps: utilizing a bulk delay unit to provide adelay substantially in accordance with a listener echo bulk delay BL;and utilizing a transversal filter unit whose coefficients are a_(k)(n), where k=0, 1, 2, . . . , M-1, where M is a number of taps in thetransversal filter unit, and whose output is substantially ##EQU4##where n is a discrete time index and g() is an output of the firstswitching unit. B_(L) is, where desired, substantially determined by aselected talker far echo ranging method.

Exemplary determinations of the listener echo estimate e_(L) (n), thelistener echo phase correction estimate Ψ(n), the first corrected signalz(n), the rotated error input η(n), the received communication signalphase correction estimate φ(n) the data phase corrected communicationsignal u(n), the rotated decision signal d(n), and related values areunderstood to be in correspondence with the determinations described forthe device as described above.

Alternatively, the method of the present invention may be selected toinclude the steps of: utilizing a noise prediction filter (NPF) receiveand filter the equalized signal, so that a noise spectrum of theequalized signal is whitened; and utilizing an inverse noise predictionfilter (NPF⁻ 1) for filtering an error signal from the second correctingunit to provide a filtered error signal for updating equalizercoefficients.

Clearly, in a preferred embodiment, the device of the present inventionmay be utilized in a modem. An exemplary modem utilizing the presentinvention is shown in FIG. 11. Such a modem typically has a talker echocancelling device, the modem being utilized for communicatinginformation with a remote device in both directions over a channel, themodem comprising at least: a transmission unit (1102), operably coupledto receive the information to be communicated, for preparing an analogsignal which carries that information for transmission; a hybridcoupling unit (1104), operably coupled to the transmission unit and to areceiving unit (1106), for transmitting the analog signal and forreceiving another analog signal from a remote device; a talker echoestimating unit (1108), operably coupled to the transmission unit(1102), for providing at least a talker echo estimate to the receivingunit (1106); and the receiving means (1106), operably coupled to thetalker echo estimating unit (1108) and to the hybrid coupling unit(1104), for recovering information from the analog signal transmitted bythe remote device.

The transmission means typically comprises at least: an encoding unit(1110), operably coupled to receive information, for providing anencoded data signal; a modulating unit (1112), operably coupled to theencoding unit (1110), for providing a modulated encoded signal; atransmission (TX) filtering unit (1114), operably coupled to the firstmodulating unit (1112), for providing a modulated, filtered transmissiondigital signal at a desired sampling rate; a digital-to-analogconverting unit (D/A) (1116), operably coupled to the transmissionfiltering unit (1114), for converting the modulated filteredtransmission digital signal to an analog signal to provide to the hybridcoupling unit (1104). The receiving unit (1106) typically comprises: ananalog-to-digital converting unit (A/D) (1118), operably coupled to thehybrid coupling unit (1104), for converting received analog signals todigital signals at a preselected sampling rate; a first combining unit(1120), operably coupled to the talker echo estimating unit (1108) andto the A/D unit (1118), for combining the received digital signal andthe talker echo estimate to provide a first talker echo correctedsignal; a rate converting unit (1122), where desired, operably coupledto the first combining unit (1120), for converting the first talker echocorrected signal, being sampled at a transmitter sampling clock, to asecond talker echo corrected signal, being sampled at a desired receiversampling clock; a demodulation unit (DEMOD)(1124), operably coupled tothe rate converting unit (1122) for providing a demodulated secondtalker echo corrected signal; a receiving (RX) filtering unit (1126),operably coupled to the demodulation unit (1124), to receive thedemodulated signal, for filtering the demodulated second talker echocorrected signal; an equalizing unit (1128), operably coupled to thereceiving filtering unit (1126), for providing an equalized,demodulated, filtered, second talker echo corrected signal; a listenerecho cancelling unit LEC (1130), operably coupled to the equalizing unit(1128), for providing a listener echo corrected, equalized, demodulated,filtered, second talker echo corrected signal; and a decoding unit(1132), operably coupled to the listener echo correcting unit (1142),for decoding the listener echo corrected, equalized, demodulated,filtered, second talker echo corrected signal to recover, substantially,the received information transmitted by the remote modem.

Typically, in the embodiment illustrated in FIG. 11, the talker echoestimating unit (1108) includes at least: a near echo correcting unit(NEC)(1134), operably coupled to the modulator (1112) and to the firstcombining unit (1120), for providing a near echo estimate as is known inthe art; a bulk delaying unit (BD)(1136), operably coupled to themodulator (1112), for providing a desired bulk delay; a far echocorrecting unit (FEC) (1138), typically including a phase-correctingunit to compensate a far echo frequency offset, operably coupled to thebulk delaying means (1136), for providing a phase-corrected far echoestimate as is known in the art; a second combining unit (1140),operably coupled to the near echo correcting unit (1134) and to the farecho correcting unit (1138), for combining the near echo estimate andthe phase-corrected far echo estimate to provide at least the firsttalker echo estimate to the first combining unit (1120).

In the embodiment illustrated in FIG. 11, the listener echo correctingunit (1130) typically comprises at least: a first correcting unit,operably coupled to the equalizing unit and to a second correcting unit, and being configured for at least: receiving the equalized,demodulated, filtered, second talker echo corrected signal; determiningat least a listener echo estimate; and combining the equalized,demodulated, filtered, talker echo corrected signal with the listenerecho estimate to provide at least a first corrected signal; wherein thesecond correcting means is operably coupled to the first correctingmeans and to the equalizing means, and is configured substantially forat least: providing at least a phase correction to the first correctedsignal to generate and output substantially a second corrected signalthat is an equalized, listener echo-corrected, and phase-correctedcommunication signal; providing at least an error signal to theequalizing means for adjusting an equalization process of the equalizingmeans, and to the first correcting means for adjusting the listener echoestimate; providing at least a rotated reference signal inreference-directed training mode for facilitating an initial training ina listener echo estimation. The listener echo cancelling unit of themodem is configured substantially as described above, and functionssubstantially as described above.

Although exemplary embodiments are described above, it will be obviousto those skilled in the art that many alternations and modifications maybe made without departing from the invention. Accordingly, it isintended that all such alterations and modifications be included withinthe spirit and scope of the invention as defined in the appended claims.

We claim:
 1. A digital communication system (DCS) comprising:(A) acommunication signal transceiver, and (B) a listener echo canceller(LECR), operably coupled to the communication signal transceiver, forreducing a listener echo interference in the digital communicationsystem having a selected sampling interval, and having an input and anoutput, said input comprising a sum of a desired signal and a listenerecho interference, and said output being free of listener echointerference, said listener echo canceller comprising:a bulk delay linemeans, operably coupled to receive and store at least one sample of oneof: said input and said output of said listener echo canceller, toprovide a bulk delay line output wherein the bulk delay line output issubstantially determined by a preselected talker far echo delaymeasuring method; an adaptive listener echo interference estimatormeans, operably coupled to receive the bulk delay line output forutilizing a listener echo bulk delay relative to the received datasignal in place of a talker far echo bulk delay relative to the nearecho to generate an estimate of the listener echo interference; and aninterference cancelling means, operably coupled to receive said input ofthe listener echo canceller and the estimate of the listener echointerference, for producing said output by subtracting said estimate ofthe listener echo interference from said input of the listener echocanceller.
 2. The DCS of claim 1, wherein said adaptive listener echointerference estimator means further includes:a set of filtercoefficients; and a tapped delay line, operably coupled to receive andstore samples of said bulk delay line output; and wherein the saidestimate of the listener echo interference is related to a weightedcombination of the stored samples of said bulk delay line output suchthat the weighting is determined by the filter coefficients, and saidfilter coefficients are adaptively adjusted to produce said estimate ofthe listener echo interference.
 3. The DCS of claim 1, wherein the saidbulk delay line output is one of:said input of the LECR, delayed by aselected number of sampling intervals; and said output of the LECR,delayed by a selected number of sampling intervals;wherein said selectednumber of sampling intervals is non-zero.
 4. The DCS of claim 1, whereinsaid adaptive listener echo interference estimator means furtherincludes compensating means, operably coupled to receive said input ofthe LECR and to receive a phase-corrected listener echo estimate from alistener echo phase correction unit, for compensating a frequency offsetin said estimate of the listener echo interference.
 5. The DCS of claim4, wherein the means for compensating the frequency offset includes:aphase-error estimating means, operably coupled to receive said input ofthe LECR and to receive a phase-corrected listener echo estimate, forgenerating a phase error that is a phase difference between saidphase-corrected listener echo estimate and said input of the LECR; and aphase-locked loop, operably coupled to receive the said phase error, forgenerating a phase correction estimate; and a phase-correction means,operably coupled to receive the phase correction estimate and forgenerating said estimate of the listener echo interference.
 6. The DCSof claim 1 wherein the communication signal transceiver is a part ofdata terminal equipment (DTE) of the DCS.
 7. Data terminal equipment(DTE) in a data communication system, comprising:(A) a communicationsignal transceiver, and (B) a device, operably coupled to thecommunication signal transceiver, for minimizing a listener echo signalinterference in a selectively demodulated communication signal receivedover a channel having simultaneous transmission and reception, thedevice comprising:equalizing means, operably coupled to receive thedemodulated, where desired, communication signal, for providing anequalized signal; a first correcting means, operably coupled to theequalizing means and to a second correcting means, and being configuredfor:receiving the equalized signal; determining a listener echoestimate; and combining the equalized signal with said listener echoestimate to provide a first corrected signal; wherein the secondcorrecting means is operably coupled to the first correcting means andto the equalizing means, and is configured for:providing a phasecorrection to the first corrected signal to generate and output a secondcorrected signal that is an equalized listener echo corrected andphase-corrected communication signal; providing an error signal to theequalizing means for adjusting an equalization process of the equalizingmeans and to the first correcting means for adjusting the listener echoestimation; and providing a rotated reference signal inreference-directed training mode for facilitating an initial training ofa listener echo estimation.
 8. The DTE of claim 7, wherein the firstcorrecting means of the device comprises:(A) primary adding means,operably coupled to the equalizing means and to a listener echoestimating means together with a listener echo phase correcting meansthat provide a phase-corrected listener echo estimate, for obtaining adifference between the equalized signal and that listener echo estimate,being the first corrected signal; (B) listener echo estimating means,operably coupled to one of:the primary adding means in a linear feedbackconfiguration, and the equalizing means in a linear feedforwardconfiguration; and further being operably coupled to the secondcorrecting means, for utilizing:in a data-receiving mode, one of:thefirst corrected signal, and the equalized signal; and in areference-directed training mode, the rotated reference signal; todetermine a first listener echo estimate; (C) listener echo phase errordetermining means, operably coupled to the listener echo phasecorrecting means and one of:in the data-receiving mode, the preliminarydecision means, and in the reference-directed training mode, a referencesignal means, for determining a listener echo phase error; wherein thelistener echo phase-locked loop means is operably coupled to thelistener echo phase error determining means for utilizing the listenerecho phase error to provide a listener echo phase correction estimate;and wherein the listener echo phase correction inverse means is operablycoupled to the listener echo phase-locked loop means and to the secondcorrecting means that provides an error signal, for utilizing thelistener echo phase correction estimate and the second correcting meanserror signal for providing a rotated error signal for updating thelistener echo estimating means.
 9. The DTE of claim 8, further includingthat the second correcting means of the device comprises:(A) receiveddata phase correcting means, operably coupled to the first correctingmeans and to a received data phase-locked loop means, for determining adata phase correction and utilizing that data phase correction toprovide a data phase corrected communication signal; (B) preliminarydecision means, operably coupled to the received data phase correctingmeans, for providing a preliminary estimate of the communication signaltransmitted by a remote modem; (C) received data phase error determiningmeans, operably coupled to the received data phase correcting means andto one of:in the data-receiving mode, the preliminary decision means,and in the reference-directed training mode, the reference signal means,for determining a received data phase error α; (D) received dataphase-locked loop means, operably coupled to the received data phasecorrecting means and to a received data phase correction inverse means,for providing the received data phase correction estimate φ(n);thereceived data phase correction inverse means being operably coupled tothe received data phase-locked loop means and to one of:the preliminarydecision means, and the reference signal means, to provide a rotateddecision signal; (E) a combiner, operably coupled to the received dataphase correction inverse means and to the first correcting means, forsubtracting the first corrected signal from the rotated decision signalto provide an error signal ε(n).
 10. The DTE of claim 8, furtherincluding that the listener echo estimating means of the deviceincludes:a bulk delay line unit, operably coupled to one of:in thelinear feedback configuration, the primary adding means, and in a linearfeedforward configuration, the equalizing means, and further beingoperably coupled to the second correcting means, for providing a delayin accordance with a listener echo bulk delay B_(L) ; and an adaptivetransversal filter unit, operably coupled to bulk delay line unit, whosecoefficients are a_(k) (n), where k=0, 1, 2, . . . , M-1, where M is anumber of taps in the transversal filter unit, and whose output is of aform: ##EQU5## where n is a discrete time index and g() is input to thebulk delay line unit.
 11. The DTE of claim 10, wherein, for the device,B_(L) is determined by a talker far echo ranging method.
 12. The DTE ofclaim 9, further including that, for the device, the phase-correctedlistener echo estimate is of a form: e_(L) (n)=p(n)e^(j)Ψ(n), where n isa discrete time index, p(n) is the first listener echo estimate, andΨ(n) is a listener echo phase correction estimate.
 13. The DTE of claim12, further including that, for the device, Ψ(n) is determined utilizingone of:(A) a listener echo phase error θ(n) where ##EQU6## where e_(Lr)and e_(Li) are the real and imaginary parts, respectively, of e_(L) (n),and q_(r) and q_(i) are the real and imaginary parts, respectively, ofthe equalized signal q(n); and further including that a listener echophase-locked loop of the first correcting unit determines an estimate ofthe listener echo frequency offset component f_(L) (n) and a phasecorrection estimate Ψ(n) for correcting the phase of the signal p(n),wherein:

    f.sub.L (n+1)=f.sub.L (n)+b.sub.1 θ(n),

and

    Ψ(n+1)=Ψ(n)+b.sub.2 θ(n)+b.sub.3 f.sub.L (n+1),

where b₁, b₂, and b₃ are constants that determine a phase-locked loopcharacteristic, and f_(L) (n) may be initialized using a talker far echofrequency offset estimate to facilitate an initial convergence; and (B)a sign of θ(n), denoted as s(n):

    s(n)=sign[e.sub.Li (n)q.sub.r (n)-e.sub.Lr (n)q.sub.i (n)],

defining

    s(n)=1 for e.sub.Li (n)q.sub.r (n)-e.sub.Lr (n)q.sub.i (n)>TH,

    s(n)=-1 for e.sub.Li (n)q.sub.r (n)-e.sub.Lr (n)q.sub.i (n)<TH,

and

    s(n)=0

otherwise, where TH is a small predetermined positive threshold value;and determining

    f.sub.L (n+1)=f.sub.L (n)+d.sub.1 s(n),

and

    Ψ(n+1)=Ψ(n)+d.sub.2 s(n)+b.sub.3 f.sub.L (n+1),

where d₁ and d₂ are selected constants and f_(L) (n), an estimate of thelistener echo frequency offset component, may be initialized using atalker far echo frequency offset estimate to facilitate an initialconvergence.
 14. The DTE of claim 12, wherein, for the device, the firstcorrected signal, z(n), is of a form:

    z(n)=q(n)-e.sub.L (n),

where q(n) is the equalized signal.
 15. The DTE of claim 14, furtherincluding that, for the device, the rotated error input, η(n) is of aform:

    η(n)=ε(n)e.sup.-jΨ(n),

where ε(n) is the error signal from the second correcting means.
 16. TheDTE of claim 15, further including that, for the device, φ(n) isdetermined utilizing a received communication signal data phase error αwhere: ##EQU7## which is used to estimate a frequency offset of areceived communication signal f_(D) (n) and the received data phasecorrection estimate φ(n), such that:

    f.sub.D (n+1)=f.sub.D (n)+c.sub.1 α(n);

and

    φ(n+1)=φ(n)+c.sub.2 α(n)+c.sub.3 f.sub.D (n+1),

where u(n) is the first corrected signal further phase-corrected byφ(n), u_(r) and u_(i) are the real and imaginary parts, respectively, ofu(n), v_(r) and v_(i) are the real and imaginary parts, respectively, ofv(n), and c₁, c₂, and c₃ are constants that determine a desiredphase-locked loop characteristic for a phase-locked loop having an inputα(n).
 17. The DTE of claim 16, further including that, for the device,the second corrected signal, u(n), is of a form:

    u(n)=z(n)e.sup.jφ(n),

n being the discrete time index and z(n) being the first correctedsignal.
 18. The DTE of claim 17, further including that, for the device,a rotated decision signal d(n) is of a form:

    d(n)=v(n)e.sup.-jφ(n),

n being the discrete time index, and v(n) being the reference signalr(n) for selection of a reference directed training mode and being apreliminary decision output for selection of a communication signalreceiving mode.
 19. The DTE of claim 9, further including, for thedevice:a noise prediction filter (NPF), operably coupled between theequalizing means and the first correcting means to receive and filterthe equalized signal, so that a noise spectrum of the equalized signalis whitened; and an inverse noise prediction filter (NPF⁻¹), operablycoupled between the second correcting means and the equalizing means,for filtering the error signal from the second correcting means toprovide a filtered error signal for updating equalizer coefficients.